Having deferred the work on the “micro multibander” for some time I finished another small QRP rig (this one for 7MHz) that is suitable for my summer excursions by bike or hiking the local mountains here in the State of Rhineland-Palatinate or the Black Forest that is not that far away on the other side of the Rhine valley.
Besides, this transceiver to be discussed here is some sort of a “remake” of a 20 meter rig I built 3 years before. And this time, the transceiver really fits into a shirt pocket without having to wear “XXXXL”- clothing. ;-):
General circuit description (instead of presenting a block diagram)
The rig uses two mixers NE602 plus one filter as central elements. The signal way is reversed when switching from receive to trasmit mode. This is done by 2 relays and is a well known technique for simple QRP rigs. You will find lots of equivalent ideas on the internet (Example 1, Example 2).
But not to ignore the shortcomings of these designs: They are somehow inferior to my requirements, particularly concercing receiver performance. I prefer to have higher signal gain and an AGC circuit. AGC for me is a must. But these designs can be expanded easily, so I added an AGC controlled interfrequency amplifier with dual gate MOSFET BF998 into the receiver’s signal path enhancing performance significantly.
The frequency generation of the superhet transceiver scheme is simple: Again I use one interfrequency (i. e. 9MHz). The VFO is DDS based on AD9835 operating below the desired radio frequency, which means that it is set to the range of about 2 MHz. Due to this low frequency you could replace the DDS by a VFO if you don’t like the relatively complex work with the software programming and microcontroller stuff). A 2MHz VFO can also be made very stable, so this is an alternative not to be ignoered.
Due to the fact that the schematic is not very difficult to analyze you are kindly requested to refer to it for further talking:
In the center of the schematic you can see the main elements of the circuit: One SSB filter (9MHz), correctly terminated by 2 resistors of 1k each (to ensure proper filter response curve) and two relays with a double set of switches. These relays reverse the way the signal travels through the filter. The advantage of this: You can use the integrated oscillator of the NE612 controlled by a crystal and a tuning capacitor to set the carrier frequency correctly for the lower sideband because the mixer is used as SSB generator and as product detector in common.
A word on chosing the proper relays: An intense examination of the relays’ data sheet is essential. I built a prototype of this transceiver on a breadboard prior to soldering the components to a veroboard. I found that some SMD relays have signifikant coupling capacities between the unused relay contacts (in the range of some Picofarads). So stray coupling was a severe problem. Later I used some second-hand Teledyne RF relays that I had purchased via ebay two years ago (price originally 50€!) for 1€ each. These relays are absolutely superb!
Before we go: In the circuit scheme above I missed out the antenna switch relay because I think every homebrewer knows what to do in this case. 😉 So the receiver’s signal path starts with a band filter for 7MHz consisting of to tuned LC circuits. The coupling is relatively loose. As coils I use the well known coil formers in TOKO style with 5.5mm outside measure.
Coil data for the 7MHz band pass filter (BPF) is 39 turns primary and 9 turns secondary of 0.1 mm enameled wire. The respective capacitor is 33pF. This is a high L to C ratio which gives you excellent LC quality factor. This is mandatory especially when working on the 40 meter band, because of the strong broadcasters starting from 7.200 kHz intermodulation might be a problem when the receiver is connected to a high gain antenna and broadcasters’ signals might overload the first mixer (remember that NE612 has a relatively low IM3!). If you still should have problems coping with too strong out-of-band signals you can reduce the coupler from 4.7pF down to 2.7pF.
In practical terms I could not detect any unwanted signal products even when using an antenna with high rf output voltage. One reasons for this is, that there is no rf preamplifier for the receiver. This avoids overloading the first mixer generally.
The NE612 has two mixer inputs and two outputs. This makes it very suitable for this sort of radio. In receive mode pin 2 of the right NE612 is used as signal input. VFO signal is fed into pin 6. The resulting mixer products are taken out from pin 4. Next the 9MHz filter follows from right to left.
The 9MHz IF signal then is fed into an IF amplifier. This one is equipped with a dual gate MOSFET (BF998), gain is about 15dB when full AGC voltage is applied wich leads to about 6V by the 1:1 volatge divider in the applied to gate 2 of the MOSFET.
The left NE612 is the product detector. I use the internal oscillator with a 9MHz crsytal and a tuning capacitor here. This saves building an extra oscillator and simplifies the rig again.
One AF low pass filter made of 1k resistor, 100uF rf choke and a 0.1 uF capacitor eliminates high frequency remainders generated by the mixing process.
The audio stages are also made simple: One preamplifier (using bipolar transistor in grounded emmitter circuit) and a final stage with LM386 transform the signal to a level that is sufficient to be fed into a small 8 ohm loudspeaker or a set of standrd MP3-player headphones. Because the rig is very small and there was definetely no space for a loudspeaker I use headphones instead.
Keep an eye on the power supply switching of the two audio stages. The problem was to eliminate the switching click and pops to a minimum and to avoid acoustic feedback when unsing a loudspeaker. So the audio preamp is only connected to DC on receive. When switching to transmit the charged capacitors avoid instant cut off supplying some Milliseconds DC to the amp until significantly discharged. The main amplfier on the other hand is connected to permanent DC supply. So it won’t pop when switching from tx to rx an vice versa but can cause feedback. To avoid feedback a transistor is used to cut the speaker/earphone from the power amplifier.
AGC is audio derived. A two stage amplifier provides a DC voltage analog to the audio input sginal strength. First amplifier stage is a common emitter bipolar transistor supplying sufficient audio voltage. This voltage is rectified by a two diode circuit letting only the positive halfways pass. You can use silicon diodes (1N1418) oder Schottky diodes here. An electrolytic capacitor (100uF/6V) provides the time constant respectively DC decay once the signal has disappeared. Output of the DC stage is split. The collector is connected to 12V via a 4.7k resistors causing a voltage drop when the transitor’s conductivity increases. The emitter is fed to the ADC of the microcontroller (pin ADC1) causing a proportional voltage to the voltage of the applied audio signal so that on the OLED an S-meter can be displayed.
An electret microphone picks the operator’s voice. The signal output level of these microphones is high enough to drive the left NE612 (which serves as balanced modulator in this case) directly. Signal input for the mixer should be 200mV RMS according to data sheet. An electret produces about 0.5 to 1 V pp if spoken with a decent voice in the distance of some centimeters. So you have more than enough audio signal power for the modulator.
BTW: Carrier suppression of the modulator is excellent. I achieved 56dB without doing anything else!
The resulting DSB signal then is fed into the SSB filter, the SSB signal subsequently is directly sent into the right NE612. A band pass filter for 7 MHz eliminates the unwanted mixer products. You should have 400 to 500 mV pp of rf signal here when the transmitter input is fully driven. I recommend a two-tone test generator to check out the linearity of this and the remaining amplifier stages!
Next parts of the transmitter are a band pass filter (same coils and capacitors like th rx bandpass filter), a preamplifier and a driver. The later should put out about 150 mW into a 50 ohm load. They are made more linear by emitter degeneration (4.7 and 2.2 ohm resistors for predriver and driver) and negative feedback. This helps to ensure that transmitter performance is fine when IMD3 products are concerned even if the main IMD3 problems usually occur in the final stage.
To transfer the rf power into the final stage proper impedance matching is mandatory. Input impedance of the final stage is fairly low (<10ohms), therefore a broadband (down)transformer is used. Data is: Core T37-43, primary 12 turns, secondary 4 turns of 0.4 mm enamled wire.
Last stage is a single ended linear amplifier in AB mode equipped with a 2SC1969 rf power transistor by eleflow.com.
BIAS circuit: The combination of the 1k resistor, a silicon diode (1N4002 or equ.) and a capacitor sets up the correct bias. Bias is fed into the cold end of the input transformer. Quiescant current should be around 40mA. A good thermal contact between the diode and the transistor is recommended. As the transistor gets warmer the diode will increase its conductivity so reducing bias current. This will prevent thermal runaway effectively!
To avoid bulky output transformers the PI-filter (7MHz LPF) is part of the tank circuit of the final amplifier transistor. For this power level this is an acceptable and practical solution because the output impedance of the stage is nearly equivalent to 50 Ohms. A certain mismatch is not a severe problem. DC to the final transistor is applied via an rf choke, for exact data please refer to the schematic!
T2 helps to suppress unwanted signals that I encountered when taking the transmitter from the dummy load test environment to a real antenna. I observed unwanted parasetic oscillation in the range of about 1MHz. T2 has a low reactance for this frequency range thus eliminating the oscillations in a reilable way by short circuiting them towards ground.
Powered with 12.5V DC the transmitter will put out slightly more than 5 watts PEP.
AD9835 is a simple but well performing 10-bit DDS chip made by Analog Devices (AD). It is controlled via 3 SPI lines transmitting the frequency data. Maximum output frequency is around 16MHz when the chip is clocked with its maximum clock rate of 50 MHz. Oscillator output voltage is some hundred millivolts peak-to-peak, so you can connect the output directly to pin 6 of the NE612 mixer.
Control signals come from an Arduino Pro Mini board. The microcontroller in this module is, if you are an Arduino user, preinstalled with a bootloader program. I overwrote this small portion of code and use the ATMega168, which is the core of the Arduino, in “native” mode. My software is written in C and transferred via “AVR dude” software using the ISP lines MOSI, MISO, SCK and RESET. These lines are not in the schematic, please refor to ATmega168 data sheet. Alternatively you can use, like shown in the schematic, an ATmega168 controller. So you have to de neccessary wiring on your own.
You will find the source code here. I packed it into an Open Document Text File because of problems I encountered when I tried to store the code into this Blogtext. If you need a compiled HEX-file, please feel free to email me!
Display is a very small OLED with 64 by 32 pixels. The OLED is, to my point of view, a little bit noisy. To suppress any rf traveling on VDD line I use an 82 ohm resistor and a set of bypass capacitors of 100uF and 0.1uF capacity closely connected to the OLED VDD pin to GND.
A low pass filter by the output of the DDS ensures spectral purity and avoids clock oscillator feed through. Remember that if you need another output frequency other than 2 MHz you should redesign the low pass filter.
Tuning is done by a rotary encoder connected to PD5 and PD6 of the microcontroller. I use the pull up resistors internal to the microcontroller, so you won’t see any other things than the mere encoder.
Tunings steps are selected by pushing the encoder knob or another suitable push button. This button is connected to ADC0 in the ATMega168 via a 3.9k resistor. The resulting ADC voltage might be problem because of a certain variation in the values of the pull up resistors that form the second resistor of the voltage divider. There is an outcommented section in the code that will show you the exact ADC value that has to be typed into the code so that key recognition works exactly.
The button once pushed will increase the tuning step by a certain amount of Hz. Steps are 10, 50, 100 (standard step), 250, 500, 1000 and 5000 Hz in and endlessly revolving chain. The step will be reset to 100Hz (standard tuning step) by leaving the tuning knob idle for 2 seconds. That’s all with the controls. Very simple, but sufficient.
The transceiver is constructed on a double sided veroboard with 6 by 8 centimeters area. Components are through hole and SMD where available. The Arduino is mounted to the front panel (another Veroboard carrying the controls etc.) as well as the OLED is. The veroboard is inserted into an aluminium frame connected to the front panel with 4 lateral M2 screws:
Wiring can be made by using the colored lines stripped from old parallel printer cables. These cables have a diameter of precisely 1mm an fit through the holes of the veroboard excactly.
If you connect any external components that are not on the same veroboard use standard 2.54 mm (0.1″) male and female board connectors! This will make it much easier to dismantle and reassemble the rig in case troubleshooting is neccessary.
Use M2 srews instaed of M3 when building very small rigs like this one!
The reverse side of the main arrangement:
Two brass made bends (from the local hardware store and each cut to a length of 8 centimeters) hold the PCB inside the mounting frame. A winding has been cut into the brass to fix the bends with screws in M2.
Together with 2 halves of a bent aluminium cabinet covered with “DC-fix” (a German manufacturer of self-adhesive PVC coating) the final rig looks like that:
So, that’s the end of the story so far. Now it’s time for going outdoor and test the rig in field use. 😉
73 and thanks for watching!